參數(shù)資料
型號(hào)： ISL6532CR-T
廠(chǎng)商： INTERSIL CORP
元件分類(lèi)： 穩(wěn)壓器
英文描述： PWM Control Circuit 16-PDIP 0 to 70
中文描述： SWITCHING CONTROLLER, 280 kHz SWITCHING FREQ-MAX, PQCC20
封裝： 6 X 6 MM, PLASTIC, MO-220VJJB, QFN-20
文件頁(yè)數(shù)： 11/14頁(yè)
文件大?。?/td> 365K
代理商： ISL6532CR-T
11
the modulator is simply the input voltage (V
IN
) divided by the
peak-to-peak oscillator voltage
V
OSC
.
Modulator Break Frequency Equations
The compensation network consists of the error amplifier
(internal to the ISL6532) and the impedance networks Z
IN
and Z
FB
. The goal of the compensation network is to provide
a closed loop transfer function with the highest 0dB crossing
frequency (f
0dB
) and adequate phase margin. Phase margin
is the difference between the closed loop phase at f
0dB
and
180 degrees. The equations below relate the compensation
network’s poles, zeros and gain to the components (R
1
, R
2
,
R
3
, C
1
, C
2
, and C
3
) in Figure 5. Use these guidelines for
locating the poles and zeros of the compensation network:
1. Pick Gain (R
2
/R
1
) for desired converter bandwidth.
2. Place 1
ST
Zero Below Filter’s Double Pole (~75% F
LC
).
3. Place 2
ND
Zero at Filter’s Double Pole.
4. Place 1
ST
Pole at the ESR Zero.
5. Place 2
ND
Pole at Half the Switching Frequency.
6. Check Gain against Error Amplifier’s Open-Loop Gain.
7. Estimate Phase Margin - Repeat if Necessary.
Compensation Break Frequency Equations
Figure 6 shows an asymptotic plot of the DC-DC converter’s
gain vs. frequency. The actual Modulator Gain has a high
gain peak due to the high Q factor of the output filter and is
not shown in Figure 6. Using the above guidelines should
give a Compensation Gain similar to the curve plotted. The
open loop error amplifier gain bounds the compensation
gain. Check the compensation gain at FP2 with the
capabilities of the error amplifier. The Closed Loop Gain is
constructed on the graph of Figure 6 by adding the
Modulator Gain (in dB) to the Compensation Gain (in dB).
This is equivalent to multiplying the modulator transfer
function to the compensation transfer function and plotting
the gain.
The compensation gain uses external impedance networks
Z
FB
and Z
IN
to provide a stable, high bandwidth (BW) overall
loop. A stable control loop has a gain crossing with
-20dB/decade slope and a phase margin greater than 45
degrees. Include worst case component variations when
determining phase margin.
Output Voltage Selection
The output voltage of the VDDQ PWM converter can be
programmed to any level between VIN and the internal
reference, 0.8V. An external resistor divider is used to scale
the output voltage relative to the reference voltage and feed
it back to the inverting input of the error amplifier, see
Figure 6.
However, since the value of R1 affects the values of the rest of
the compensation components, it is advisable to keep its
value less than 5kW. Depending on the value chosen for R1,
R4 can be calculated based on the following equation:
0.8V
×
DDQ
If the output voltage desired is 0.8V, simply route V
DDQ
back
to the FB pin through R1, but do not populate R4.
The output voltage for the internal V
TT
linear regulator is set
internal to the ISL6532 to track the V
DDQ
voltage by 50%.
There is no need for external programming resistors.
Component Selection Guidelines
Output Capacitor Selection - PWM Buck Converter
An output capacitor is required to filter the inductor current
and supply the load transient current. The filtering
requirements are a function of the switching frequency and
the ripple current. The load transient requirements are a
function of the slew rate (di/dt) and the magnitude of the
transient load current. These requirements are generally met
with a mix of capacitors and careful layout.
DDR memory systems are capable of producing transient
load rates above 1A/ns. High frequency capacitors initially
supply the transient and slow the current load rate seen by the
bulk capacitors. The bulk filter capacitor values are generally
determined by the ESR (Effective Series Resistance) and
voltage rating requirements rather than actual capacitance
requirements.
FLC
2
π
x
LO
x
CO
------------------------------------------
=
FESR
x
ESR
x
CO
-------------------------------------------
=
F
Z1
R
2
x C
2
------ x
=
F
Z2
R
1
R
3
(
)
x
C
3
------ x
=
F
P1
2
π
x
R
2
x
C
2
1
C
2
+
---------------------
---------------------------------------- x
=
F
P2
R
3
x
C
3
------ x
=
100
80
60
40
20
0
-20
-40
-60
F
P1
F
Z2
10M
1M
100K
10K
1K
100
10
OPEN LOOP
ERROR AMP GAIN
F
Z1
F
P2
20LOG
(R
2
/R
1
)
F
LC
F
ESR
COMPENSATION
GAIN
CLOSED LOOP
GAIN
G
FREQUENCY (Hz)
20LOG
(V
IN
/
V
OSC
)
MODULATOR
GAIN
FIGURE 6. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN
R4
0.8V
–
V
=
ISL6532
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