參數(shù)資料
型號(hào)： ISL6532CCRZ-T
廠商： INTERSIL CORP
元件分類： 穩(wěn)壓器
英文描述： R8C Series, 21 Group, WDTO 48P6Q-A
中文描述： SWITCHING CONTROLLER, 280 kHz SWITCHING FREQ-MAX, PQCC28
封裝： 6 X 6 MM, LEAD FREE, PLASTIC, MO-220VJJC, QFN-28
文件頁(yè)數(shù)： 14/16頁(yè)
文件大?。?/td> 506K
代理商： ISL6532CCRZ-T
14
converter’s ripple current and the ripple voltage is a function
of the ripple current. The ripple voltage and current are
approximated by the following equations:
Increasing the value of inductance reduces the ripple current
and voltage. However, the large inductance values reduce
the converter’s response time to a load transient.
One of the parameters limiting the converter’s response to a
load transient is the time required to change the inductor
current. Given a sufficiently fast control loop design, the
ISL6532C will provide either 0% or 100% duty cycle in
response to a load transient. The response time is the time
required to slew the inductor current from an initial current
value to the transient current level. During this interval the
difference between the inductor current and the transient
current level must be supplied by the output capacitor.
Minimizing the response time can minimize the output
capacitance required.
The response time to a transient is different for the
application of load and the removal of load. The following
equations give the approximate response time interval for
application and removal of a transient load:
where: I
TRAN
is the transient load current step, t
RISE
is the
response time to the application of load, and t
FALL
is the
response time to the removal of load. The worst case
response time can be either at the application or removal of
load. Be sure to check both of these equations at the
minimum and maximum output levels for the worst case
response time.
Input Capacitor Selection - PWM Buck Converter
Use a mix of input bypass capacitors to control the voltage
overshoot across the MOSFETs. Use small ceramic
capacitors for high frequency decoupling and bulk capacitors
to supply the current needed each time the upper MOSFET
turns on. Place the small ceramic capacitors physically close
to the MOSFETs and between the drain of upper MOSFET
and the source of lower MOSFET.
The important parameters for the bulk input capacitance are
the voltage rating and the RMS current rating. For reliable
operation, select bulk capacitors with voltage and current
ratings above the maximum input voltage and largest RMS
current required by the circuit. Their voltage rating should be
at least 1.25 times greater than the maximum input voltage,
while a voltage rating of 1.5 times is a conservative
guideline. For most cases, the RMS current rating
requirement for the input capacitor of a buck regulator is
approximately 1/2 the DC load current.
The maximum RMS current required by the regulator may be
closely approximated through the following equation:
For a through hole design, several electrolytic capacitors
may be needed. For surface mount designs, solid tantalum
capacitors can be used, but caution must be exercised with
regard to the capacitor surge current rating. These
capacitors must be capable of handling the surge-current at
power-up. Some capacitor series available from reputable
manufacturers are surge current tested.
MOSFET Selection - PWM Buck Converter
The ISL6532C requires 2 N-Channel power MOSFETs for
switching power and a third MOSFET to block backfeed from
V
DDQ
to the Input in S3 Mode. These should be selected
based upon r
DS(ON)
, gate supply requirements, and thermal
management requirements.
In high-current applications, the MOSFET power dissipation,
package selection and heatsink are the dominant design factors.
The power dissipation includes two loss components;
conduction loss and switching loss. The conduction losses are
the largest component of power dissipation for both the upper
and the lower MOSFETs. These losses are distributed between
the two MOSFETs according to duty factor. The switching
losses seen when sourcing current will be different from the
switching losses seen when sinking current. When sourcing
current, the upper MOSFET realizes most of the switching
losses. The lower switch realizes most of the switching losses
when the converter is sinking current (see the equations below).
These equations assume linear voltage-current transitions and
do not adequately model power loss due the reverse-recovery of
the upper and lower MOSFET’s body diode. The gate-charge
losses are dissipated in part by the ISL6532C and do not
significantly heat the MOSFETs. However, large gate-charge
increases the switching interval, t
SW
which increases the
MOSFET switching losses. Ensure that both MOSFETs are
within their maximum junction temperature at high ambient
temperature by calculating the temperature rise according to
package thermal-resistance specifications. A separate heatsink
may be necessary depending upon MOSFET power, package
type, ambient temperature and air flow.
I =
V
IN
- V
OUT
Fs x L
V
OUT
V
IN
V
OUT
=
I x ESR
x
t
RISE
=
L x I
TRAN
V
IN
- V
OUT
t
FALL
=
L x I
TRAN
V
OUT
I
RMSMAX
V
IN
-------------
I
OUTMAX
2
1
12
------
V
----------------------------
V
s
–
V
IN
-------------
×
2
×
+
×
=
P
LOWER
= Io
2
x r
DS(ON)
x (1 - D)
Where: D is the duty cycle = V
OUT
/ V
IN
,
t
SW
is the combined switch ON and OFF time, and
f
s
is the switching frequency.
Approximate Losses while Sourcing current
Io
2
r
DS ON
Approximate Losses while Sinking current
P
UPPER
= Io
2
x r
DS(ON)
x D
P
LOWER
Io
2
r
DS ON
)
×
1
D
–
(
)
×
1
2
--
Io
V
IN
×
t
SW
f
s
×
×
+
=
P
UPPER
)
×
D
×
1
2
--
Io
V
IN
×
t
SW
f
s
×
×
+
=
ISL6532C
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