
REF19x Series 
Rev. I | Page 20 of 28 
ON
OFF
10k
1k
5%
NC
NC
OUTPUT
V
IN
NC
NC
REF19x
NC = NO CONNECT
8
7
6
5
1
2
3
4
1μF
TANT
+
0
Figure 22. Membrane Switch-Controlled Power Supply 
CURRENT-BOOSTED REFERENCES WITH 
CURRENT LIMITING 
While the 30 mA rated output current of the REF19x series is 
higher than is typical of other reference ICs, it can be boosted to 
higher levels, if desired, with the addition of a simple external 
PNP transistor, as shown in Figure 23. Full-time current 
limiting is used to protect the pass transistor against shorts. 
U1
REF196
(SEE TABLE)
R4
2
R1
1k
R2
1.5k
Q2
2N3906
+
C2
100μF
25V
D1
R3
1.82k
C1
10μF/25V
(TANTALUM)
S
F
C3
0.1μF
F
S
R1
Q1
TIP32A
(SEE TEXT)
+V
 = 6V
TO 9V
(SEE TEXT)
V
COMMON
V
C
V
COMMON
OUTPUT TABLE
U1
REF192
REF193
REF196
REF194
REF195
V
OUT
 (V)
2.5
3.0
3.3
4.5
5.0
+V
OUT
3.3V
@ 150mA
2
6
1N4148
(SEE TEXT
ON SLEEP)
3
+
4
0
Figure 23. Boosted 3.3 V Referenced with Current Limiting 
In this circuit, the power supply current of reference U1 flowing 
through R1 to R2 develops a base drive for Q1, whose collector 
provides the bulk of the output current. With a typical gain of 
100 in Q1 for 100 mA to 200 mA loads, U1 is never required to 
furnish more than a few mA, so this factor minimizes tempera-
ture-related drift. Short-circuit protection is provided by Q2, 
which clamps the drive to Q1 at about 300 mA of load current, 
with values as shown in Figure 23. With this separation of 
control and power functions, dc stability is optimum, allowing 
most advantageous use of premium grade REF19x devices for 
U1. Of course, load management should still be exercised. A 
short, heavy, low dc resistance (DCR) conductor should be used 
from U1 to U6 to the V
OUT
 Sense Point S, where the collector of 
Q1 connects to the load, Point F. 
Because of the current limiting configuration, the dropout 
voltage circuit is raised about 1.1 V over that of the REF19x 
devices, due to the V
BE
 of Q1 and the drop across Current Sense 
Resistor R4. However, overall dropout is typically still low 
enough to allow operation of a 5 V to 3.3 V regulator/reference 
using the REF196 for U1 as noted, with a V
S
 as low as 4.5 V and 
a load current of 150 mA. 
The requirement for a heat sink on Q1 depends on the maximum 
input voltage and short-circuit current. With V
S
 = 5 V and a 300 
mA current limit, the worst-case dissipation of Q1 is 1.5 W less 
than the TO-220 package 2 W limit. However, if smaller TO-39 
or TO-5 packaged devices, such as the 2N4033, are used, the 
current limit should be reduced to keep maximum dissipation 
below the package rating. This is accomplished by simply 
raising R4. 
A tantalum output capacitor is used at C1 for its low equivalent 
series resistance (ESR), and the higher value is required for 
stability. Capacitor C2 provides input bypassing and can be an 
ordinary electrolytic. 
Shutdown control of the booster stage is an option, and when 
used, some cautions are needed. Due to the additional active 
devices in the V
S
 line to U1, a direct drive to Pin 3 does not 
work as with an unbuffered REF19x device. To enable shutdown 
control, the connection from U1 to U2 is broken at the X, and 
Diode D1 then allows a CMOS control source, V
C
, to drive U1 
to U3 for on/off operation. Startup from shutdown is not as 
clean under heavy load as it is in basic REF19x series, and can 
require several milliseconds under load. Nevertheless, it is still 
effective and can fully control 150 mA loads. When shutdown 
control is used, heavy capacitive loads should be minimized. 
NEGATIVE PRECISION REFERENCE WITHOUT 
PRECISION RESISTORS 
In many current-output CMOS DAC applications where the 
output signal voltage must be the same polarity as the reference 
voltage, it is often necessary to reconfigure a current-switching 
DAC into a voltage-switching DAC using a 1.25 V reference, an 
op amp, and a pair of resistors. Using a current-switching DAC 
directly requires an additional operational amplifier at the 
output to reinvert the signal. A negative voltage reference is 
then desirable, because an additional operational amplifier is 
not required for either reinversion (current-switching mode) or 
amplification (voltage-switching mode) of the DAC output 
voltage. In general, any positive voltage reference can be 
converted into a negative voltage reference using an operational 
amplifier and a pair of matched resistors in an inverting 
configuration. The disadvantage to this approach is that the 
largest single source of error in the circuit is the relative 
matching of the resistors used. 
The circuit illustrated in Figure 24 avoids the need for tightly 
matched resistors by using an active integrator circuit. In this 
circuit, the output of the voltage reference provides the input 
drive for the integrator. To maintain circuit equilibrium, the 
integrator adjusts its output to establish the proper relationship 
between the reference’s V
OUT
 and GND. Thus, any desired 
negative output voltage can be selected by substituting for the 
appropriate reference IC. The sleep feature is maintained in the 
circuit with the simple addition of a PNP transistor and a 10 kΩ 
resistor.